Mobile electronic communication devices—including cellular telephones, pagers, smartphones, remote monitoring and reporting devices, and the like—have dramatically proliferated with the advance of the state of the art in wireless communication networks. Many such devices are powered by one or more batteries, which provide a Direct Current (DC) voltage. One challenge to powering electronic communication devices from batteries is that the battery does not output a stable DC voltage over its useful life (or discharge cycle), but rather the DC voltage decreases until the battery is replaced or recharged. Also, many electronic communication devices include circuits that operate at different voltages. For example, the Radio Frequency (RF) circuits of a device may require power supplied at a different DC voltage than digital processing circuits.
A DC-DC converter is an electrical circuit typically employed to convert an unpredictable battery voltage to one or more continuous, regulated, predetermined DC voltage levels, and thus to provide stable operating power to electronic circuits. Numerous types of DC-DC converters are known in the art. The term “buck” converter is used to describe a DC-DC converter that outputs a lower voltage than the DC source (such as a battery); a “boost” converter, also called a step-up, is one that outputs a higher voltage than its DC input. Both boost and buck converters may be implemented as switched mode power supplies (SMPS), in which energy is transferred from a source, such as battery, to a storage component, such as an inductor or capacitor, at a high frequency through transistor switches.
Supplying power to an RF power amplifier (PA) of an electronic communication device is particularly challenging. The efficiency of an RF PA varies with the amplitude of the transmission signal to be amplified. Maximum efficiency is achieved at full power, and drops rapidly as the transmission signal amplitude decreases, due to transistor losses accounting for a higher percentage of the total power consumed. The loss of efficiency may be compensated by a technique known as “envelope tracking,” in which the output of a DC-DC converter, and hence the voltage supplied to the PA, is not constant, but is modulated to follow the amplitude modulation of the transmission signal. In this manner, at any given moment, the power supplied to the RF PA depends on the amplitude of the signal being amplified. Such modulation of the RF PA power supply can dramatically improve power consumption efficiency.
FIG. 1 depicts the relevant RF output portion of an electronic communication device 10. A battery 12 provides a battery voltage VBAT to an efficient, wide-bandwidth envelope-tracking power supply 14 that modulates the supply voltage of the RF PA 16. The RF PA 16 outputs an amplified RF signal for transmission from the device 10 on one or more antenna 18. The modulated voltage VCC(t) output by the dynamic power supply 14 should be capable of tracking a rapidly varying reference voltage. As such, the power supply 14 must meet certain bandwidth specifications. The required bandwidth depends on the specifications of the network(s) in which the RF PA 16 is used. For example, the required bandwidth exceeds 1 MHz for EDGE systems (8PSK modulation), and exceeds 30 MHz for LTE20 (Long Term Evolution).
The design of transmitters 10 employing envelope tracking power supplies 14 for RF PAs 16 is challenging, and requires the use of more sophisticated characterization techniques than is the case for designing traditional, fixed supply power amplifiers. The fundamental output characteristics of an RF PA 16 with an envelope tracking power supply—power, efficiency, gain, and phase—depend on two control inputs: RF input signal power and the supply voltage VCC.
A typical envelope tracking system dynamically adjusts the supply voltage VCC to track the RF input signal envelope at high instantaneous power. In this case, the PA 16 operates with high efficiency in compression. The instantaneous supply voltage VCC(t) primarily determines the PA 16 output characteristics. Unfortunately, this supply voltage modulation introduces an additional source of distortion that is due to the variations of PA 16 gain and/or phase as a function of the supply voltage VCC. FIG. 2A depicts variations in gain (AM-AM), and FIG. 2B depicts variations in phase AM-PM) with supply voltage VCC variations. In a conventional PA, supplied by a constant voltage, these variations are only linked to the variation of the RF input signal voltage.
FIG. 3A depicts the time-domain variations of the voltage VCC supplied to a PA 16 by an envelope tracing power supply 14, and FIG. 3B depicts the corresponding time-domain variations in the gain of the PA 16. These waveforms clearly show the dependency of the gain on supply voltage modulation. For example, large negative deviations, or drops, in supply voltage VCC correspond to very large decreases in gain. Moreover, this dependency is a nonlinear one, being significantly more pronounced as the supply voltage VCC is reduced. FIG. 4 depicts the PA 16 gain as a function of the supply voltage VCC, where again it is evident that low excursions of supply voltage VCC correspond to dramatic drops in the small signal gain of the RF PA 16. This modulation of the PA 16 gain impairs linearity of the PA 16.
In general, the transfer function for an amplified RF signal isVOUT(VCC,VIN)=GPA(VCC,VIN)·VIN  (1)where VIN is the magnitude of the RF PA 16 input signal, VOUT(VIN,VCC) is the magnitude of the output signal, VCC is the PA 16 instantaneous voltage from power supply 14, and finally GPA(VIN,VCC) is the gain of the PA. The gain GPA, and hence the output voltage VOUT, is a nonlinear function of VIN and VCC. Ideally, the gain should be constant.
The nonlinear gain GPA can be approximated by a two-dimensional polynomial.GPA(VCC,VIN)=b0+b1VCC+b2VCC2+ . . . +bβ-1VCCβ-1+bβVCCβ
with:
                                          b            0                    =                                    c                              0                1                                      +                                          c                                  1                  1                                            ·                              V                IN                                      +            …            +                                          c                                  α                  -                                      1                    1                                                              ·                              V                IN                                  α                  -                  1                                                      +                                          c                                  α                  1                                            ·                              V                IN                α                                                    ⁢                                  ⁢                              b            1                    =                                    c                              0                2                                      +                                          c                                  1                  2                                            ·                              V                IN                                      +            …            +                                          c                                  α                  -                                      1                    2                                                              ·                              V                IN                                  α                  -                  1                                                      +                                          c                                  α                  2                                            ·                              V                IN                α                                                    ⁢                                  ⁢        …        ⁢                                  ⁢                              b                          β              -              1                                =                                                    c                                  0                                      β                    -                    1                                                              ·                              V                IN                                      +            …            +                                          c                                  α                  -                                      1                                          β                      -                      1                                                                                  ·                              V                IIN                                  α                  -                  1                                                      +                                          c                                  α                                      β                    -                    1                                                              ·                              V                IN                α                                                    ⁢                                  ⁢                              b            β                    =                                    c                              0                β                                      +                                          c                                  1                  β                                            ·                              V                IN                                      +            …            +                                          c                                  α                  -                                      1                    β                                                              ·                              V                IN                                  α                  -                  1                                                      +                                          c                                  α                  β                                            ·                              V                IN                α                                                                        (        2        )            
FIGS. 5A and 5B depict how laboratory measurements (circles) of PA 16 gain vs. supply voltage VCC and RF transmission signal voltage, respectively, can be interpolated using the 2-D polynomial (continuous lines). Note that the phase shift of the PA 16 could similarly be written this way and interpolated by a 2-D polynomial.
The term “ISO-Gain” is used herein as a generic, descriptive term to denote a constant gain (GISO) in an RF PA 16, which does not change in response to variations in VCC and VIN. From equation (1), the gain can be written as:
                                          G            PA                    ⁡                      (                                          V                CC                            ,                              V                                  I                  ⁢                                                                          ⁢                  N                                                      )                          =                                                            V                OUT                            ⁡                              (                                                      V                    CC                                    ,                                      V                    IN                                                  )                                                    V              IN                                =                                                                      V                  OUT                                ⁡                                  (                                                            V                      CC                                        ,                                          V                      IN                                                        )                                                            V                CC                                      ·                                          V                CC                                            V                IN                                                                        (        3        )            
In envelope tracking operation the voltage VCC (from the voltage supply 14) is a linear replica of the envelope of an RF input signal multiplied by a constant α. As a result:
                                          G            PA                    ⁡                      (                                          V                CC                            ,                              V                IN                                      )                          =                                                            V                OUT                            ⁡                              (                                                      V                    CC                                    ,                                      V                    IN                                                  )                                                    V              CC                                ·          α                                    (        4        )            
In order to obtain a constant gain GISO over variations in both VCC and VIN, the following condition must be met:
                                                        G              PA                        ⁡                          (                                                V                  CC                                ,                                  V                  IN                                            )                                =                      G            ISO                          ⁢                                  ⁢                  so          ,                                    (        5        )                                          G          ISO                =                                                            V                OUT                            ⁡                              (                                                      V                    CC                                    ,                                      V                    IN                                                  )                                                    V                              CC                ⁢                                                                  ⁢                _                ⁢                                                                  ⁢                pred                                              ·                      α            .                                              (        6        )            
This condition can be fulfilled by modifying the shape of VCC through a particular pre-distortion gain that depends on both VCC and VIN:VCC—pred(VCC,VIN)=VCC·Gainpred(VCC,VIN).  (7)
Substituting equation (7) into equation (6) yields an expression of the required pre-distortion gain:
                                          Gain            pred                    ⁡                      (                                          V                CC                            ,                              V                IN                                      )                          =                              α                          G              ISO                                ·                                                    V                OUT                            ⁡                              (                                                      V                    CC                                    ,                                      V                    IN                                                  )                                                    V              CC                                                          (        8        )            FIG. 6 depicts the operation of ISO-Gain pre-distortion in the case where VCC is constant and VIN (or PIN) varies. The nonlinear gain applied to the supply voltage VCC is depicted in the lower curve, whereas the PA 16 gains with and without ISO-Gain pre-distortion are depicted in the upper two curves. The upper curve clearly shows that ISO-Gain operation allows maintaining the gain of the PA 16 relatively constant for a wide range of VIN (or PIN).
In practice, pre-distortion such as that depicted in FIG. 6 relies on testing and calibrations performed at chip fabrication. Pre-distortion parameters are calculated, and stored in large look-up tables (LUT). During operation, pre-distortion perturbations are retrieved from the LUTs and applied to the power supply 14. Accuracy is proportional to the size of the LUT; however, accuracy is limited because the necessary computation time is also proportional to the LUT size. Additionally, because all known envelope tracking RF PA power supply pre-distortion is “open-loop” and uses factory-generated values, the pre-distortion applied cannot be recalculated over the life of the circuit, such as to account for shifts in RF frequency, output power, temperature, component aging, and the like.
The Background section of this document is provided to place embodiments of the present invention in technological and operational context, to assist those of skill in the art in understanding their scope and utility. Unless explicitly identified as such, no statement herein is admitted to be prior art merely by its inclusion in the Background section.